Power supply

ABSTRACT

A switching regulator for regulating an ac signal, and method of supplying power thereto wherein the regulator includes a positive half cycle part and a negative half-cycle part arranged to regulate the respective parts of an input ac signal. Each half-cycle part includes a modulating transistor, having an associated modulator diode, and a clamping diode arranged to protect the modulating transistor from reverse-bias voltages and having an associated clamp switch. The regulator further has a first switching controller operable to cause the modulating transistors to switch and a second, separate switching controller operable to cause the clamp switches to switch.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is a National Phase Application of InternationalApplication No. PCT/GB2006/004952, filed Dec. 28, 2006, which claimspriority to Great Britain Patent Application No. 0526625.9 filed Dec.30, 2005, which applications are incorporated herein fully by thisreference.

FIELD OF THE INVENTION

The present invention relates to a power supply that receives power froman ac source and provides at least two output signals. The outputsignals may be supplied, for example, to a switching regulator thatregulates an ac signal. The switching regulator may find application in,for example, a domestic combined heat and power (dchp) generator thatproduces an ac signal that may require regulating prior to supply toeither connected appliances or to an electrical grid.

BACKGROUND OF THE INVENTION

There are many situations where a generated ac signal requiresregulating before it can be used elsewhere. The regulation may controlfrequency, voltage or current and may include control of voltage andcurrent transients and steady state variations that may otherwise causefluctuations in the ac waveform to be used elsewhere.

To provide a context for the present invention an intended applicationfor the present invention will be described. This application is in dchpunits that provide hot water and central heating in a domesticenvironment. Our International Patent Application No. PCT/GB03/001200describes such a dchp unit comprising a Stirling engine. These dchpunits are beneficial as, in addition to meeting a household's centralheating and hot water requirements, they can also be used to generateelectricity in an energy-efficient manner. The electricity so generatedcan be used either within the household or it may be sold back into theelectrical grid supplying the household.

The electricity generated by the dchp unit must be tightly regulated tobe suitable for supply onto a mains electrical grid. In addition,regulation is also usually required to suit the demands of domesticappliances connected to the dchp unit.

FIG. 1 shows the well-known Buck regulator that may be used to regulatea DC waveform. The regulator comprises a transistor that is switched,normally according to a pulse width modulation scheme, to provide adesired average output voltage. The inductor and capacitor smooth thepulsed output to leave only a minimal ripple on the DC voltage signalprovided at the output. The regulator also includes a diode provided toact as a clamping diode (also known as a “flyback diode”) to protect thetransistor from large reverse voltages generated by the inductor as ittries to maintain current flow when the transistor is switched off.

A pair of Buck regulators may be combined to provide a regulator for anac supply. FIG. 2 reproduces such a regulator that is disclosed inEP-A-0,631,372. The regulator converts an ac input to provide a variablevoltage dc output to power lights operated from a dimmer switch. A pairof transistors with associated diodes are provided, as indicated at A inFIG. 2, with one transistor modulating the positive half-cycle of the acinput and the other transistor modulating the negative half-cycle of theac input. A pair of clamping diodes are provided, as indicated at B inFIG. 2, one for each half-cycle and biased appropriately, that areswitched in and out of the circuit for the appropriate half-cycles byassociated transistors. Thus, both positive and negative half-cycles arepulse width modulated, and the resulting output is smoothed by theinductors and capacitors.

SUMMARY OF THE INVENTION

Against this background, and from a first aspect, the present inventionresides in a switching regulator for regulating an ac signal. Theregulator is provided with input terminals for receiving an ac inputsignal and output terminals for providing an output signal. An inductorand a capacitor are provided such that they operate to smooth the signalpassing through the regulator thereby providing a smoothed output signalat the output terminals. The regulator operates separately on thepositive and negative half-cycles of the ac input signal and so isprovided with a positive half-cycle part and a negative half-cycle part.

Each of the positive and negative half-cycle parts include the followingcomponents. A modulating transistor is provided that is operable tomodulate its respective half-cycle of the input ac signal. Themodulating transistor is provided with an associated modulator diodethat is arranged to allow current flow through the modulating transistorduring that modulating transistor's respective half-cycle and to resistcurrent flow through the modulating transistor during the otherhalf-cycle. Thus, the modulator diode ensures that the modulatingtransistor may operate on the correct respective half-cycle of the acinput signal. Generally, the modulator diodes of the two half-cycleparts will be biased oppositely.

Each of the positive and negative half-cycle parts further comprises aclamping diode arranged to protect the modulating transistor fromreverse-bias voltages. The clamping diode is provided with an associatedclamp switch operable to connect the clamping diode into the regulatorduring that modulating transistor's respective half-cycle and todisconnect the clamping diode from the regulator during the otherhalf-cycle. Thus, the clamping diode is only active during the requiredhalf-cycle.

The regulator further comprises a first switching controller operable tocause the modulating transistors to switch and a second switchingcontroller operable to cause the clamp switches to switch. The first andsecond switching controllers are separate. Accordingly, control of themodulating transistors may be effected independently of control of theclamp switches.

Preferably, the regulator operates using pulse width modulation of theac input signal to provide the required output signal. This may beachieved by arranging the first switching controller to cause themodulating transistors to switch in accordance with a pulse widthmodulation scheme.

The modulator diodes may be arranged to provide a shunt around theirassociated modulating transistor as this provides a simple arrangementto bypass each of the respective modulating transistors during thehalf-cycle where the modulating transistor is not required to work.

Transistors are preferably used as the clamp switches. Optionally, theclamp transistors are arranged on respective shunts. Thus, when theclamp transistors are conducting, their associated clamping diode isbypassed such that the clamping diode may be switched out of theregulator during the half-cycle for which it would be incorrectlybiased.

Preferably, the first switching controller includes a power supply thatis operable to take power from the input ac signal. A correspondingarrangement may be used for the second switching controller, eitheralone or in combination with the power supply for the first switchingcontroller. When used in combination, a first power supply may beprovided for the first switching controller that is arranged to takepower from one half-cycle of the input ac signal. A second power supplymay be provided for the second switching controller that is arranged totake power from the other half-cycle of the input ac signal. This may beachieved by providing oppositely-biased diodes arranged to allow onlyeither the positive or the negative half-cycle of the input ac signal toflow to the first and second power supplies respectively.

Preferably, the first and second power supplies are arranged to obtainsubstantially equal average quantities of power from the input acsignal. Thus, the power requirement across the whole ac cycle of theinput signal is balanced and so the integrity of the input signal isonly minimally affected, if at all. The modulating transistors and theclamp switches may have very different current requirements, for examplebecause of different switching frequencies. As a consequence, the firstand second power supplies may be arranged to provide substantiallydifferent instantaneous currents. This means that it is likely the powersupplies will have to decrease the voltage of the signal drawn from theinput ac signal, and so they may be arranged to do this. For example,one or both the power supplies may comprise a switch operable todecrease the voltage. Alternatively, the voltage may be capped at adesired voltage using an appropriately-biased Zener diode. Preferably,one or both power supplies are operable to provide a smoothed powersignal. This may be conveniently achieved by providing one or both powersupplies with smoothing components such as an inductor and/or acapacitor. Furthermore, one or more capacitors may be provided in one orboth power supplies that are arranged to store energy during onehalf-cycle and to release energy during the other half-cycle. In orderto provide a desired current, one or both power supplies may be providedwith appropriately-rated resistors.

From a second aspect, the present invention resides in a method ofproviding power to any of the above described switching regulators,comprising powering the first switching controller by drawing power fromone half-cycle of the input ac signal; and powering the second switchingcontroller by drawing power from the other half-cycle of the input acsignal; wherein the average power drawn during each half-cycle issubstantially the same. The method may further comprise using the firstswitching controller to provide current to the modulating transistors,and using the second switching controller to provide current to theclamp switches, wherein the instantaneous currents provided by the firstand second switching controllers are substantially different.

BRIEF DESCRIPTION OF THE DRAWINGS

In order that the present invention may be more readily understood,preferred embodiments will now be described, by way of example only,with reference to the accompanying drawings in which:

FIG. 1 is a circuit diagram of known Buck regulator;

FIG. 2 is a circuit diagram of a known ac regulator that essentiallycombines two Buck regulators;

FIG. 3 is a block diagram of a power supply comprising first and secondpower supply units;

FIG. 4 is a block diagram of a regulator operable to receive an ac inputsignal and produce an ac output signal;

FIG. 5 is a circuit diagram of part of an ac regulator according to thepresent invention;

FIG. 6 is a circuit diagram of an ac regulator according to the presentinvention, including the circuit of FIG. 5 and also showing associatedswitching controllers;

FIG. 7 is a circuit diagram of a driver circuit for supplying first andsecond gate drive signals;

FIG. 8 is a circuit diagram of a clamp switching controller;

FIG. 9 is a circuit diagram of a modulator switching controller;

FIG. 10 is a circuit diagram for a power supply of one of the switchingcontrollers of FIG. 6 that controls a pair of modulating transistors;

FIG. 11 is a circuit diagram for a power supply of one of the switchingcontrollers of FIG. 6 that controls a pair of transistors associatedwith clamping diodes;

FIG. 12 is a graph showing an uneven ac input signal and offsets usedwhen changing from positive to negative switching and back again;

FIG. 13 a is a block diagram of a method of regulating an ac signalaccording to an embodiment of the present invention;

FIG. 13 b is a schematic diagram of an implementation of the method ofFIG. 13 a.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 3 shows a power supply 10 that draws power from an input ac signal12. The input ac signal 12 is also provided to a pair of outputterminals 14 to which other components or circuits may be connected. Thepower supply 10 draws power from live and neutral lines 16,18 comprisingthe input ac signal 12 via lines 20,22. These lines 20,22 divide so asto provide power to a first power supply unit (PSU) 24 and a second PSU26.

The first and second PSU's 24,26 are configured to draw power duringopposed halves of the input ac signal 12, i.e. first PSU 24 draws powerduring the positive half-cycles of the input ac signal 12 and the secondPSU 26 draws power during the negative half-cycles or vice versa. Thismay easily be implemented using appropriately biased diodes, forexample.

The first and second PSU's 24,26 are configured to ensure that power isdrawn evenly from the positive half-cycles as compared to the negativehalf-cycles. This minimises the effect seen on the input ac signal 12appearing at the output terminal 14, such as any distortion. Furthermorethe first and second PSU's 24,26 are configured to modify the signalthey receive so as to provide output signals having differentinstantaneous currents to their respective output terminals 28,30. Thesignals the first and second PSU's provide may be dc or may be ac, asrequired. Further components or circuitry may be connected to the PSUoutput terminals 28,30 that may be separate or part of circuitryconnected to output terminals 14.

Specific examples of the PSU's 24,26 are provided in the descriptionbelow, along with a specific application where the power supply 10 maybe used.

Regulator Assembly

FIG. 4 shows, in block form, a regulator assembly 101 that includesparts suitable for implementing the present invention. As can be seen at96, an ac input signal is received on live and neutral lines 104,106that are passed to an ac regulator 100. The ac input signal may befloating, i.e. it may not necessarily be referenced to ground. The acregulator 100 performs the actual regulation of the ac input signal andis operated under the control of the other parts of the regulatorassembly 101 shown in FIG. 4.

As will become clearer from the following description, the ac regulator100 comprises four transistors; two switching to modulate the ac signalthereby regulating the ac input signal, and two that switch to allowclamp diodes to become effective to protect the modulating transistors.The clamp transistors operate as controlled by a clamp switchingcontroller 126 that is powered by a clamp power supply unit (PSU) 142that in turn draws its power from the ac input signal.

Similarly, the modulating transistors have an associated modulatorswitching controller 124 powered by a modulator PSU 124 that draws powerfrom the ac input signal.

The modulator switching controller 124 merely switches operation betweena pair of modulating transistors to allow one or the other modulatingtransistor to operate. The actual switching of each modulatingtransistor is controlled according to a pulse width modulation (PWM)scheme. Accordingly, a PWM module 90 receives a signal from themodulator switching controller 124 that will allow one or the other (orneither) modulating transistor to operate. The PWM module 90 alsoreceives signals from a voltage comparator 92 and a current comparator94.

The voltage comparator 92 references the ac input signal and operates toeffect modulation of the ac input signal to control the voltage of theac output signal. The current comparator 94 also references the ac inputsignal, but also receives a signal 99 that is indicative of the currentflowing through one or more loads connected to the regulator assembly101. The current comparator 94 operates to effect modulation of the acinput signal to control the current provided to the load.

The regulator assembly 101 may operate in one of two modes: a voltagecontrol mode under the management of the voltage comparator 92, or acurrent control mode under the management of the current comparator 94.The current drawn by the load may be monitored to determine which modeis used. Where normal currents are required, the voltage control modemay be used, i.e. the voltage comparator 92 controls the PWM module 90that in turn controls the ac regulator 100. However, where excessivecurrents are required, the current control mode may be used (at leastfor short periods), i.e. the current comparator 94 controls the PWMmodule 90 that in turn controls the ac regulator 100. This is describedin more detail in the following section.

In any event, there may be a limit provided by a current overloaddetector 97 that also receives the signal 99 that is indicative of thecurrent flowing through one or more loads connected to the regulatorassembly 101. If the current becomes too large, or an excessive currentpersists for too long, this may be interpreted as a short circuit andthe current overload detector may operate to stop operation of the acregulator 100. This may be achieved by the current overload detector 97controlling the PWM module 90.

The regulator assembly 101 described above may be used to connect agenerator such as a dchp generator to a grid or to connected appliances.For example, the regulator assembly 101 may form a bridge between thealternator of a dchp unit and an electrical grid and also localelectrical appliances to ensure that the signal produced by thealternator is suitable for injection into the grid and/or supply to theconnected appliances.

Voltage Control and Current Control

As mentioned above, a contemplated application for the present inventionis in a Stirling engine in a dchp unit that produces an ac signal fromits alternator. A particular requirement for such a low-inertiagenerators is to provide a suitable impedance across the generatorterminals, irrespective of load demand. If the alternator senses toohigh or too low an impedance, this may result in over-voltage, inwaveform distortion and, in extreme cases such as an open or shortcircuit condition, in physical damage to the generator.

The alternator is assured of being presented with a reasonably stableimpedance when it is connected directly to an electrical mains.Furthermore, the alternator is protected against damaging faults andtransients by monitoring circuits that are fitted in accordance withregulatory requirements. However, there is no inherent protection forthe alternator when such a dchp unit is used to provide electricalenergy to connected appliances when disconnected from the electricalmains, as in the case of a grid power blackout. Under these conditions,the load corresponding to the connected appliances that is connectedacross the electrical output of the alternator may vary from nothing upto the full rated output of the alternator. In fact, when appliances arefirst connected to the dchp unit, these loads may demand “inrush”currents greatly in excess of those normally provided by the alternator.

It is advantageous to ensure that such a low inertia generator ispresented with a stable impedance under all load demand conditions, andthis is implemented using the voltage and current control modes ofoperation. Also, the current control mode of operation provides amechanism for accommodating the inrush current of an electricalappliance at first connection as described below.

In this embodiment, the ac input signal will have a nominal voltage anda maximum current. For example, the ac input signal may be produced bythe alternator in a dchp unit operating to produce a 230V rms signalwith a maximum current of 4.3 A.

The current drawn by the connected loads may be monitored to determinewhether or not it exceeds that 4.3 A limit. While a current of 4.3 A orless is drawn, the regulator may operate in voltage control mode suchthat the waveform of the ac output signal is tightly controlled tofollow the ideal sinusoid having an amplitude of 230V rms. In this mode,excess current may be dumped to a dump resistor such that the alternatorsees a constant impedance.

However, in many situations a current of more than 4.3 A may berequired. For example, if a toaster is connected to the regulator, itwill demand a large current when first switched on. The cold heatingelements may draw as much as 24 A initially. This large current demandmay be sensed as a voltage drop as a capacitor in the ac regulator 100discharges (and so is indicative of the current drawn), and so theregulator assembly 101 may change to current control mode.

In current control mode, a constant current is drawn from the alternatorat the 4.3 A maximum. This power is supplied to an inductor in the acregulator 100. The ac output signal derived form the inductor may have avoltage that is allowed to drop below 230V rms to ensure that thecurrent rises above 4.3 A to meet the demand using the available power.Thus, the voltage is allowed to vary while the current comparator 94operates to control the current drawn from the alternator at the maximumvalue while allowing higher currents to be delivered to the connectedloads.

As with many situations, the toaster will only draw a large current fora short time period, while the heating elements are warming. When hot,the toaster will only require typically 2.4 A, well within the usualoperating range of the regulator. Thus, operation may switch back tovoltage control mode once the current demand drops below the maximumvalue of 4.3 A.

In fact, two different thresholds may be used to provide hysteresis thatprevents hunting (i.e. repeated switching resulting from noise in thesignal causing repeated crossings of the threshold). When operating involtage control mode, a drop to 220V rms may be used to indicate avoltage drop large enough to indicate an excessive current demand and socause a switch to current control mode. When operating in currentcontrol mode, a rise to 225V rms may be used to indicate that currentdemand is normal once more and to cause the switch to voltage controlmode. Thus, when the voltage drops through the 220V rms threshold, noisefluctuations in the signal will be too small to cross the 225V rmsthreshold so that control is not inadvertently switched back to voltagemode prematurely. The 5V difference between thresholds is chosen as itis larger than expected noise variations. Similarly, an increase involtage through the 225V rms threshold to change operation to voltagecontrol requires a subsequent large drop to 220V rms before control isswitched back to current mode, this drop again being too great to bebridged by noise in the signal.

The ac Regulator

An ac regulator 100 with which the present invention may be used isshown in FIG. 5. The ac regulator 100 comprises a pair of inputterminals 102 for connection to the ac source. In this embodiment, theinput terminals 102 receive the output of an alternator of a Stirlingengine operating in a dchp unit. The ac regulator 100 receives a nominal240V ac signal as an input between live and neutral lines 104 and 106respectively. The ac regulator 100 provides the desired ac output signalat a pair of output terminals 108. In this embodiment, the ac input fromthe dchp unit is regulated and subsequently distributed from the outputterminals 108 to a number of connected domestic appliances that drawpower from the dchp unit. In addition, the regulator 100 may provide theregulated ac signal for supply into an electrical mains supply.

Essentially, the ac regulator 100 comprises a combination of two Buckregulators. Accordingly, the ac regulator 100 comprises a pair ofmodulating transistors 110 a,b that both operate to pulse width modulatethe ac input signal to provide a desired signal as the ac output signal.Suitable PWM schemes and their implementation are well known in the art.The PWM may be performed to control the voltage or current of the acoutput signal, as described elsewhere in this specification.

One of the transistors 110 a modulates during the positive half-cycle ofthe ac input signal and the other transistor 110 b modulates during thenegative half-cycle. To allow this method of operation, the transistors110 a,b are arranged in series and each transistor 110 a,b has anassociated shunt provided with a modulator diode 112 a,b. The twomodulator diodes 112 a,b are oppositely biased such that modulatingtransistor 110 b is bypassed for the positive half-cycle of the ac inputsignal and modulating transistor 110 a is bypassed for the negativehalf-cycle.

A pair of clamping diodes 114 a,b are also provided, biased oppositelysuch that diode 114 a may act as a clamping diode during the positivehalf-cycle and diode 114 b may act as a clamping diode during thenegative half-cycle. Switched shunts 116 a,b are provided to allow eachclamping diode 114 a,b to be bypassed during the half-cycle for which itis not required to work. Switches are provided by a pair of transistors118 a,b, herein after referred to as clamp transistors 118 a,b todistinguish them from the modulating transistors 110 a,b describedabove.

An inductor 120 and a capacitor 122 are provided to smooth the signalprovided by the modulating transistors 110 a,b thereby providing therequired output signal at the output terminals 108.

The regulator 100 may be operated as follows.

During the positive half-cycle of the ac input signal, current flowsfrom the live line 104 and is blocked by modulator diode 112 a such thatthe current must flow through modulating transistor 110 a where it isgated according to the pulse width modulating scheme. Current thenbypasses modulating transistor 110 b (that is switched off) along theshunt through modulator diode 112 b. The emergent current then flows toinductor 120 and capacitor 122 that operate to smooth the current flowseen at the output terminals 108. Clamping diode 114 a protects themodulating transistor 110 a from reverse voltages when the inductor 120tries to maintain current flow. This is because clamp transistor 118 ais switched on to bypass clamping diode 114 b, and clamp transistor 118b is switched off. This ensures the only current path is from theneutral line 106 to the live line 104 via the shunt 116 a providedthrough clamp transistor 118 a and then through clamping diode 114 a.

During the negative half-cycle of the ac input signal, current flow isaccomplished from the neutral line 106 to the live line 104 viamodulating transistor 110 b. Modulator diode 112 b blocks current flowso that current must pass through the modulating transistor 110 b whereit is gated according to the pulse width modulating scheme. Currentbypasses modulating transistor 110 a (that is switched off) along theshunt through modulator diode 112 a. Again, the inductor 120 and thecapacitor 122 operate to smooth the current flow seen at the outputterminals 108. This time, the other clamping diode 114 b protects themodulating transistor 110 b when the inductor 120 tries to maintaincurrent flow. This is because clamp transistor 118 b is switched on tobypass clamping diode 114 a, and clamp transistor 118 a is switched off.This ensures the only current path is from the live line 104 to theneutral line 106 via the shunt 116 b provided through clamp transistor118 b and then through clamping diode 114 b.

The Switching Controllers

FIG. 6 shows the regulator of FIG. 5, but also shows the modulatorswitching controller 124 associated with the modulating transistors 110a,b and the clamp switching controller 126 associated with the clamptransistors 118 a,b.

The switching controllers 124, 126 require power that may be suppliedfrom associated power supply units (PSU's) 132,142 (not shown separatelyin FIG. 6). Further details of the PSU's are provided in the followingsection.

FIG. 7 shows a general driver circuit 10 for providing a pair of gatedrive signals at output terminals 12 a,b that may be used in the acregulator 100. The gate drive signals are produced with reference to anac input signal received at a live input 14 and a neutral input 16.Logic is provided such that the gate drive signals may have either highor low states. Moreover, the driver circuit 10 is arranged so that whenthe output at terminal 12 a is high, the output at terminal 12 b, islow, and vice versa. The states of the outputs at terminals 12 a,bswitch from high to low or vice versa as the ac input signal changesfrom positive to negative and back again such that the outputs atterminals 12 a,b are never both high. Further details on exactly whenthe switching is made are provided in one of the following sections.

As can be seen from FIG. 7, the logic part of the driver circuit maycomprise a pair of NOT gates 18 a,b. In this example, NOT gate 18 a isthe master logic gate. A shunt 20 including a capacitor 22 may extendaround the NOT gates 18 a,b to improve the responsiveness of the drivercircuit. The output of NOT gate 18 a provides the output at terminal 12a, and is also passed to NOT gate 18 b where it is inverted to providethe output on terminal 12 b. Hence, the outputs appearing at terminals12 a,b are a combination of high and low, as primarily controlled by theoutput of the master logic gate, NOT gate 18 a.

The circuit 10 may also include three transistors Q1, Q2 and Q3, allarranged to provide shunts to ground 24. Transistor Q3 may be switchedbetween on and off to determine whether a current flows to the input ofNOT gate 18 a and this determines the states of the gate drive signals.Transistor Q1 may be provided to clamp the live terminal 14 to ground 24when the ac input signal is negative. Similarly, Q2 may be provided toclamp the neutral terminal 16 to ground 24 when the ac input signal ispositive.

The operation of this exemplary driver circuit 10 is as follows. Assumeas a starting point that the live terminal 14 is positive on the risingside of a positive half-cycle and the neutral terminal 16 is negative onthe falling side of the negative half-cycle. Then, the positive liveterminal 14 sees current flow to Q3 such that it is conducting. Thus,current from a DC power supply 26 flows through transistor Q3 to ground24 rather than flowing to NOT gate 18 a. Thus, the input to NOT gate 18a is low and its output is high. This output is seen at output terminal12 a such that the output at 12 a is high. The high output from NOT gate18 a becomes the input to NOT gate 18 b, such that NOT gate 18 bproduces a low output that is seen at terminal 12 b.

A feedback loop 28 passes the high output from NOT gate 18 a to the baseof transistor Q2, such that transistor Q2 is conducting. Thus Q2provides a shunt that clamps the neutral terminal 15 to ground 24. Theneutral terminal 16 is also connected to transistor Q1 that is thus offin view of the shunt through transistor Q2. Of course, transistor Q1being off ensures current from the live terminal 14 flows to transistorQ3 rather than flowing straight to ground 24.

As the polarity of the ac input signal changes, the live terminal 14goes to zero and then negative with respect to neutral terminal 16.Hence, current flow to transistor Q3 fails and it turns off. With Q3off, current from the DC supply 26 flows to NOT gate 18 a. With a highinput, NOT gate 18 a produces a low output that is seen at terminal 12a. The low output from NOT gate 18 a is inverted by NOT gate 18 b tobecome a high output at terminal 12 b. The low output from NOT gate 18 ais seen at the base of transistor Q2 via feedback loop 28, such thattransistor Q2 switches off. With the neutral terminal 16 no longerclamped to ground 24 and becoming increasingly positive, current flowsto transistor Q1 to turn it on. With Q1 conducting, the live terminal 14is clamped to ground 24.

As the polarity of the ac input signal changes again, the neutral input16 falls to zero and so transistor Q1 switches off and diode D1 protectstransistors Q1 and Q2 as neutral terminal 16 becomes negative withrespect to the positive terminal 14. Thus, live terminal 14 is no longerclamped to ground 24. As the live terminal 14 becomes positive, Q3switches on so that the input to NOT gate 18 a becomes low, and thestates seen at output terminals 12 a,b reverse. The high output from NOTgate 18 a is fed to transistor Q2 that turns on to clamp the neutralterminal 16 to ground 24.

The clamping cycles may be looked at another way. When live terminal 14is positive with respect to the neutral terminal 16 (i.e. during thepositive half-cycle of the ac input signal), then the neutral input 16is clamped to ground 24 as the reference (0V) level via transistor Q2.Similarly, when the neutral input 14 is negative with respect to thelive input 16 (the negative half-cycle), the live input 14 is clamped toground 24 as the reference level via transistor Q1. By restricting thereference level switching to a region within about one volt of zerovolts and coordinating this changeover with the transitions of the drivesignals appearing at output terminals 12 a,b, this driver protectsitself and the also the drive devices (e.g. transistors) from damage dueto reverse polarity connections.

Turning now to the specific implementation of the general circuit ofFIG. 7, FIG. 8 shows the clamp switching controller 126 in detail. Inthis example, the clamp switching controller 126 receives a +15V dcsignal from the clamp PSU 142, as will be described below. As can beseen, the clamp switching controller of FIG. 8 essentially correspondsto the driver circuit 10 of FIG. 7. Similar reference numerals are usedfor similar parts, except incremented by 200.

One difference with the circuit of FIG. 8 is that the logic isimplemented using NOR gates rather than NOT gates. To ensure NOR gates218 a,b function as NOT gates, their inputs are arranged in twowell-known configurations. For NOR gate 218 a, the second input is tiedto ground 224. For NOR gate 218 b, the same signal (the output from NORgate 218 a) is supplied to both inputs. These arrangements could beswapped, or the same arrangement could be used for both NOR gates 218a,b. The outputs from the NOR gates 218 a,b are indicated at 212 a,b.These outputs 212 a,b are no longer seen as output terminals but arepassed to further NOR gates 230 a,b whose function will be describedbelow.

As will be clear from the preceding description, during the positivehalf-cycle when the live terminal 214 is positive, output 212 a is highand output 212 b is low. At this time, transistor Q3 is on to hold theinput to NOR gate low, transistor Q2 is on to clamp the neutral terminal216 to ground 224 and transistor Q1 is off to ensure that the liveterminal 214 is not clamped to ground 224. During the negativehalf-cycle when the neutral terminal 216 is positive, output 212 a islow and output 212 b is high. At this time, transistor Q3 is off toallow the input to NOR gate to be high and transistor Q2 is off bothensuring that the neutral terminal 216 is not clamped to ground 224 andallowing Q1 to turn on to clamp the live terminal 214 to ground 224.

The additional NOR gates 230 a,b are included to ensure that the twogate drive signals appearing at output terminals 212 a,b cannot beactivated while any residual voltage exists across the inductor 120 inthe ac regulator 100. This protects the clamp transistors 118 a,b fromreverse voltages when the inductor 120 is discharging by ensuring thatthe clamp transistors 118 a,b remain inoperable during this time.

In practice this is achieved by passing the output 212 a to one input ofNOR gate 230 a whose other input is connected to the “top” end of theinductor coil 120 via coil terminal 234. Thus, a high output is onlyseen on output terminal 232 a when no current flows from the inductor120 and when the output at 212 a is also low. As output 212 a is lowduring the negative half-cycle, the gate drive signal at terminal 232 ais passed to clamp transistor 118 b such that shunt 116 b is in placeduring the negative half-cycle to ensure clamping diode 114 b is active.During the negative half-cycle output 212 b is high and so the gatedrive signal at terminal 232 b is always low. This gate drive signal issupplied to clamp transistor 118 a ensuring that it remains off and thatclamping diode 114 b remains active.

The output 212 b is passed to one input of NOR gate 230 b whose otherinput is connected to the neutral terminal 216 and so sees the voltageon the “back” end of the inductor coil 120. As a result, the gate drivesignal appearing at terminal 232 b is only high when both the inductor120 has discharged in the positive half-cycle (when the output atterminal 212 b is low). Thus, during the positive half-cycle, the highoutput at terminal 232 b switches clamp transistor 118 a on and the lowoutput at terminal 232 a ensures clamp transistor 118 b is off. Thisensures clamp diode 114 a is active throughout the positive half-cycle.

At the beginning of each half-cycle, the outputs at 212 a,b reverse butany remaining voltage across inductor 120 keeps both gate drive signalsfrom appearing at terminals 232 a,b off. Hence, both clamp transistors118 a,b remain off and hence protected until the inductor 120 is fullydischarged.

FIG. 9 shows the modulator switching controller 124 that receives a 12Vsupply from modulator PSU 132. This 5V supply may be obtained from the12V provided by modulator PSU 132 in any standard fashion (in fact, themodulator PSU 132 is used to power other components requiring 12V, sohence this arrangement). The circuit is very similar to those of FIGS. 7and 8, and so similar reference numerals will be used althoughincremented by 300 relative to FIG. 7. As the modulating transistors 110a,b are protected from reverse currents produced when the inductor 120discharges (by virtue of the clamping diodes 114 a,b), there is no needto include NOR gates or a reference to the voltage on the inductor 120.Also, the live terminal 314 and neutral terminal 316 are effectivelyreversed to ensure that the modulator switching controller 124 operates180° out of phase with respect to the clamp switching controller 126.

Accordingly, the transistor Q1 still operates to clamp the live terminal314 to ground 324, the transistor Q2 still operates to clamp the neutralterminal 316 to ground 324 and transistor Q3 operates to set the inputto NOT gate 318 a. Thus, when the neutral terminal 316 is positive withrespect to the live input 314, it is seen at the base of transistor Q3that is conducting. So, the current from modulator PSU 132 flows toground 324, ensuring that the input to NOT gate 318 a is low. This meansthat the output of NOT gate 318 a is high and this is seen at outputterminal 312 a. The high output from NOT gate 318 a is passed to NOTgate 318 b ensuring that its output is low as seen on output terminal312 b. In addition, the high output from NOT gate 318 a is passed alongfeedback loop 328 to hold transistor Q1 on. Thus, the live terminal 314is clamped to ground 324 through transistor Q1. With the live terminal314 clamped to ground, transistor Q2 is held off ensuring that theneutral terminal 316 is not clamped to ground.

When the ac input signal at live input 314 goes positive with respect tothe neutral input 316, the neutral input 316 falls to zero therebyswitching Q3 off. This sees the input to NOT gate 318 a go high,resulting in a low output at terminal 312 a and a high output atterminal 312 b. The low output from NOT gate 318 a is seen by transistorQ1 via feedback loop 328, and so transistor Q1 switches off. With Q1switched off, the live terminal 314 is no longer clamped to ground 324and its now positive-going potential sees Q2 switch on thereby clampingthe neutral terminal 316 to ground 324.

So, during positive half-cycles, the gate drive signal at terminal 312 ais low and the gate drive signal at terminal 312 b is high. Conversely,during negative half-cycles, the gate drive signal at terminal 312 a ishigh and the gate drive signal at terminal 312 b is low. Terminal 312 ais connected to modulating transistor 110 b, while terminal 312 b isconnected to modulating transistor 110 a. This ensures that modulatingtransistor 110 a may be switched during the positive half-cycle (whenterminal 312 b is high) and modulating transistor 110 b may be switchedduring the negative half-cycle (when terminal 312 a is high). Asdescribed above, the gate drive signals are not supplied directly to themodulating transistors 110 a,b, but are subject to the pulse widthmodulation by the PWM module 90 that produces the required regulatedsignal. Thus, the modulator switching controller 124 operates to controlwhen the modulating transistors 110 a,b may be switched by the PWMmodule 90 and to ensure the modulating transistors 110 a,b are switchedoff at all other times.

The above embodiment uses drive signals that have high values to drivetheir connected transistors (or whatever other device they may drive).Of course, where devices operate under inverted logic (i.e. to requirelow drive signals to activate the devices rather than high signals), theabove embodiments may be readily adapted to invert their logic outputs.

The Switching Controller PSU's

The PSU's may draw power evenly from the ac input signal, e.g. from theac signal supplied by the dchp unit. As can be seen from FIG. 6, themodulator PSU 132 may draw power from the neutral line 106 via anappropriately-biased diode 128 such that the modulator PSU 132 onlyreceives power during the negative half-cycle of the ac input signal.Conversely, the clamp PSU 142 may draw power from the live line 104 viaan appropriately-biased diode 130 such that the clamp PSU 142 onlyreceives power during the positive half-cycle of the ac input signalThus, the PSU's 132,142 may draw power during alternate half-cycles.Moreover, the PSU's 132,142 may draw power evenly to ensure theintegrity of the ac signal from the dchp unit and yet may be capable ofproviding an asymmetric current waveform to the switching controllers132,142. This obviates the need for power factor correction that wouldotherwise add complexity and expense.

FIG. 10 shows an embodiment of the modulator PSU 132. As noted above,this PSU 132 may draw power during the negative half-cycle of the acinput signal. In this case, current flow is from neutral 106 to live104. The modulating transistors 110 a,b used in this embodiment requirea maximum continuous current of 40 mA at a voltage of 12V dc. The 12Vlevel may be obtained from the 240V input by using a switch 134 operatedwith a switching ratio of 20:1. A diode 136 and smoothing components(inductor 138 and capacitors 140) may be included to ensure a smooth 12Vdc output. To supply the required average current of 40 mA, themodulator PSU 132 may draw a current of 80 mA when it operates duringthe negative half-cycle. The 20:1 switching ratio sees a current of 4 mAdrawn from the ac input because the power must remain constant(remembering voltage drops from 240V to 12V across the switch).

FIG. 11 shows the clamp PSU 142 for the clamp transistors 118 a,b. Asdescribed above, this PSU 142 may draw power during the positivehalf-cycle of the ac input signal. The clamp transistors 118 a,b requirea significantly lower current as they switch far less frequently thanthe modulating transistors 110 a,b. Specifically, the clamp transistors118 a,b require 1.8 mA maximum continuous, at 15 Vdc. As a result, aswitching circuit like that of FIG. 10 is not favoured. Instead, asimple half-wave rectifying circuit may be used that may beshunt-regulated using a Zener diode 144 rated at the required 15V. Thepower consumption of this circuit is likely to be lower than that ofFIG. 10 where the switch 134 is likely to be implemented as a fieldeffect transistor.

The Zener diode 144 may be used to limit the voltage across the outputto 15V, and the parallel capacitor 146 may be used to smooth the outputand to store energy during the positive half-cycle for discharge duringthe negative half-cycle. The required average current of 1.8 mA may beobtained by drawing a 3.6 mA current during only the positivehalf-cycle. This 3.6 mA current may be obtained from the 240V inputaccording to Ohm's Law using two 33 kΩ resistors 148 in series toprovide the necessary 66 kΩ resistance.

Where these two different arrangements of the modulator PSU 132 andclamp PSU 142 are used, they ensure significantly differentinstantaneous currents may be provided to their associated transistors110 a,b, 118 a,b, yet sill allowing power to be drawn evenly from the acinput signal.

Two specific examples of PSU's 132, 142 are provided above, althoughother PSU's may be used to power the transistors 100 a,b, 118 a,b. Forexample, both PSU's 132 and 142 may be switchers, or both may be linear.They may even share a common design. Alternatively, charge pumps may beused as PSU's to multiply or divide a voltage. A suitable example is thefour-stage Dickson charge pump. Such a charge pump does not useinductors and so does not produce large magnetic fields that mayotherwise cause interference.

Timing of Polarity Switching

The preceding sections described the switching controllers 124,126 thateffect the change between positive and negative half-cycles, along witha description of their PSU's 132,142. This section describes how theexact timing of the change from positive switching to negative switchingmay be managed. This timing should be tightly controlled in order toavoid potential problems associated with the fact that the ac inputsignal is not likely to be a perfect sinusoidal signal. FIG. 11 shows anexample of an uneven ac input signal that may be obtained, for example,from a dchp unit (albeit exaggerated for the purposes of illustration).Such an uneven signal may not make a single zero crossing when changingfrom positive to negative half-cycles and vice versa. As can be seen,noise on the signal may lead to three or more zero-crossings.

Switching should be tightly controlled around zero volts to ensuretransistors 110 a,b, 118 a,b do not switch repeatedly which would be atbest inefficient and may at worst damage the transistors 110 a,b, 118a,b. In addition, switching of transistors 110 a,b, 118 a,b must becontrolled to ensure that both transistors from either pair 110 a,b or118 a,b are not switched on at the same time. In particular, the clamptransistors 118 a,b should not be allowed to be switched on at the sametime as a short circuit would form along shunts 116 a,b from live 104 toneutral 106.

In order to avoid these problems, a switching regime may be implementedthat creates a “dead zone” around zero volts in which no switching ispermitted. To this end, a pair of offsets may be used for each polaritychange: −V₁ and −V_(switch) for positive to negative switches, and +V₁and +V_(switch) for negative to positive switches. These offsets areshown in FIG. 11.

The ±V₁ offsets creates the dead zone such that a “zero volts” conditionis met whenever the voltage between the live and neutral inputs fallswithin the narrow band between ±V₁. Any transitions between positive andnegative caused by fluctuations in the ac input signal as it crossesthrough zero volts that remain within this band are not distinguishablefrom zero as far as the circuit is concerned. To this end, the valuesfor ±V₁ may be chosen to be greater than the background signal noiselevel.

The onset of active switching of the transistors 110 a,b, 118 a,b occurswhen the voltage of the ac input signal exceeds ±V_(switch). Activeswitching is stopped once the ac input signal falls and crosses the ±V₁offsets. The offsets create a band between +V₁ & +V_(switch) (forpositive signals) and a band between −V₁ & −V_(switch) (for negativesignals) that provide hysteresis to eliminate “hunting” (the rapidrepeated enabling and inhibiting of switching that would otherwise becaused by the ac input signal fluctuating above and below a singleactuation level). These bands may be set so that they are greater thanthe anticipated noise magnitude.

Starting in a positive half-cycle, modulating transistor 110 a will beactive and switching to regulate the ac input signal according to thePWM scheme. Modulating transistor 110 b is off. Clamp transistor 118 ais on and clamp transistor 118 b is off to ensure clamping diode 114 ais effective. As the ac input signal falls towards zero volts, bothmodulating transistor 110 a and clamp transistor 118 a switch off as the+V₁ threshold is crossed. Hence, all transistors 110 a,b, 118 a,b arenow switched off in advance of the ac input signal crossing zero volts.Inductor 120 will discharge once the ac input signal crosses zero voltsand so the offsets help to ensure switching does not commence before theinductor 120 has discharged fully. As discussed above, the clampswitching controller 126 is the fail safe in this respect as the gatedrive signals are controlled by logic gates 232 a,b such that neithercan go high until the voltage across the inductor 120 falls to zero.

As the ac input signal goes increasingly negative, it crosses the firstoffset −V₁. This may cause the switching controllers 124,126 to switchfrom positive to negative mode, i.e. the states of transistors Q1, Q2and Q3 switch and the live terminals 214,314 become clamped to ground224,324. While the switching controllers 124,126 are now ready foroperation, the gate drive signals are maintained at low until the secondoffset −V_(switch) is crossed. After this crossing, clamp switchingcontroller 126 may send a high drive signal to switch clamp transistor118 b on, thereby making clamping diode 114 b effective. Subsequently,clamp switching controller 124 may set a high drive signal to allowmodulating transistor 110 b to begin switching according to the requiredpulse width modulation scheme. To ensure that clamp transistor 118 bswitches on before modulating transistor 110 b, −V_(switch) for theclamp transistor 110 b may be set closer to zero volts then −V_(switch)for the modulating transistor 110 b.

When the ac input signal starts to fall once more, switching is onlystopped once the −V₁ offset is crossed.

As will be appreciated, the reverse protocol may be used when the acinput signal switches from negative to positive. Briefly, the modulatingtransistor 110 b and the clamp transistor 118 b switch off, zero voltsis crossed, +V₁ is reached at which point switching controllers 124,126go from negative to positive (Q1, Q2 and Q3 switch, neutral 216,316 isclamped to ground 224,324), and finally +V_(switch) is reached thatfirst causes clamp transistor 118 a to switch on followed by modulatingtransistor 110 a.

FIG. 12 shows example values for ±V₁ and ±V_(switch). In practice, thesevalues may be varied. As mentioned above, different values may be usedfor the modulator switching controller 124 and for clamp switchingcontroller 126, such as different values for ±V_(switch) to ensure thatthe clamp transistors 118 a,b are on before the modulating transistors110 a,b start switching. While FIG. 12 shows the pairs of offsets ±V₁and ±V_(switch) symmetrically offset from zero volts, this need not bethe case. For example, +V₁ may have a different magnitude to −V₁ due tothe effects of instantaneous unbalanced power drains that are used bythe PSU's 132,142. As will be remembered, although the average powertaken from positive to negative half-cycles is balanced, there may bevariations in the instantaneous levels.

The exact implementation of this switching scheme using the switchingcontrollers 124,126 is dependent on selection of component values thattake into account device characteristics such as transistor and diodevoltage drops, current amplification factors and voltage switchingcharacteristics. It would be straightforward for a person skilled in theart to determine appropriate choices of components and component values,either by calculation, empirical measurement or both.

Regulation of the ac Input Signal

The method by which regulation of the ac input signal is performed bythe voltage comparator 92 or current comparator 94 in conjunction withthe PWM module 90 will now be described with reference to FIGS. 13 a and13 b. Essentially, both the voltage comparator 92 and the currentcomparator 94 operate in similar fashion. This is because although thecurrent comparator 94 operates to regulate current, it implements thisby monitoring the voltage across a resistance (i.e. it effects currentcontrol indirectly through voltage control). Thus, the followingdescription applies to both operation of the voltage comparator 92 andthe current comparator 94.

The regulator assembly 101 regulates the ac input signal to provide theac output signal. At 50, the ac input signal is sampled to produce asampled ac signal as an input waveform 52. In fact, a feedback loopprovides a reference to the output signal and this may also be sampledat 50. Consequently, the sampled ac signal reflects the ac input signaland is modified to correct for inaccuracies introduced into the outputsignal downstream.

The sampled ac signal 52 may be full-wave rectified at 54 by a full-waverectifier 56 to produce the rectified ac signal. The rectified ac signalmay then be scaled at 60 with reference to a desired output voltage,230V rms in this example, to produce a scaled ac signal 58.

In parallel, the sampled ac signal 52 may be used to generate triggerpulses 62 to coincide with the sampled ac signal 52 crossing throughzero volts. This zero crossing may be detected using software, and mayuse digital filtering to remove the effects of noise around the zerocrossing and make use of software pattern matching to improve phasesynchronisation. A computer 64 uses these trigger pulses 62 to generatea synchronised reference signal 66, as shown at 68. The reference signal66 corresponds to a sinusoid, but with only positively extending lobessuch that it is equivalent to a full-wave rectified ac signal. Thereference signal 66 is synchronised to the sampled ac signal 52 usingthe trigger pulses 62 such that the reference signal 66 touches zerovolts in synchrony with the scaled ac signal 58 touching zero volts.

This sinusoidal reference signal 66 is generated using a lookup table tosupply values to a digital to analogue converter. Only values relatingto the part of a sinusoid form 0 to π/2 radians are stored: these valuesare used in reverse for the π/2 to n radians part, and this shape isrepeated for the π to 2 π radians part.

At 70, the scaled ac signal 58 may be subtracted from the referencesignal 66 to produce an error signal 72 (i.e. instantaneous values aresubtracted from instantaneous values). To ensure that only positivevalues are obtained, an offset is introduced. For example, thissubtraction may be implemented in a difference amplifier operating witha suitable offset. Thus, the error signal 72 contains only positivevalues.

As can be seen, this error signal 72 is a function of the phase of theac input signal. This phase variation may be removed at 74 by amultiplier chip 78 that operates to divide the error signal by thereference signal 66 to provide a % error signal 78. This % error signal78 may then be used at 80 to modulate the ac input signal. As bothreference signal 66 and error signal 72 are positive, the multiplierchip 78 need only operate in one quadrant. This vastly simplifies thecost and complexity of the multiplier chip 78. Moreover, otheradvantages are obtained by using this single quadrant operation. Forexample, cross-over distortion and linearity mismatch that are inherentwhen switching between quadrants (i.e. as one input changes polarity)are avoided. Moreover, any DC offset present in the ac input signal thatmay otherwise alter the operating characteristics of reactive components(such as interference filter chokes) is applied equally to bothhalf-cycles as they are both treated as positive going. Thus the overallwaveshape with reference to zero volts is unchanged: if applied topositive and negative half-cycles. The feedback loop also compensatesfor any DC offsets introduced by the regulator itself.

The modulation may be performed according to a pulse width modulationscheme to modulate out any error in the signal. For example, the % errorsignal 78 may be compared to a ramp signal to cause pulse widths to bedefined by where the ramp signal and the % error signal 78 cross. Forexample, where the regulator is producing the required ac output signal,the reference signal will match the sampled ac signal (after rectifyingand scaling) such that a zero % error signal results. This will cause afull width pulse in the modulation such that the voltage of the ac inputsignal is unchanged. This situation is unusual as the input ac signalwill generally have a greater magnitude than that required for the acoutput signal. Hence, it is far more common for the voltage of the inputac signal to be more than that required. This is reflected in thesampled ac signal and a % error signal 78 results that causes smallerpulses. The smaller pulses modulate the input ac signal to pull down thevoltage of the ac output signal to the required level.

The magnitude of the reference signal 66 and the degree of scaling ofthe sampled ac signal 52 is chosen so that a zero % error signal 78results when the ac input signal is at a desired 230V rms. An advantageof the present invention is its flexibility: generation of the referencesignal 66 and the scaling may be varied to suit any desired outputvoltage, for example to suit the local electrical grid.

It will be evident that modifications may be made to the above. Forexample, a scaling of the ac input signal relative to the referencesignal 66 is required and so the scaling may be performed on the acinput signal, the reference signal 66 or both. Also, scaling may beperformed only after the division at 74. For example, the % error may bepassed to an amplifier for scaling.

It will be clear to the skilled person that variations may be made tothe above embodiments without necessarily departing from the scope ofthe invention that is defined by the appended claims.

The above regulator assembly 101 has been described in the context ofregulating an ac supply provided by a Stirling engine in a dchp unit foruse by domestic appliances. However, regulators according to the presentinvention may find useful application elsewhere. Essentially, theregulator assembly 101 is designed to operate downstream of any voltagesource such as a generator, a mains supply, etc.

For example, the regulator assembly 101 may be used to buffer theinterface between a Stirling engine alternator and the mains supply toprevent voltage and current transients and steady state variations thatmay otherwise cause disruptive fluctuations in the output waveform. Suchan arrangement reduces the likelihood of engine shutdowns beinginitiated by the control system, as a safety precaution, in response toloss of quality of the grid supply.

A welcome advantage of the regulator assembly 101 according to thepresent invention is that it allows connection between a stand-alonegenerator and its loads (such as connected appliances) in any country ormarket whatever its electrical grid constraints. By providing a means toset the voltages for a range of possible grid and engine frequencies,the switching regulator can provide suitable voltage/frequency modelsfor use in equipment matched to different markets and their grids.

The regulator assembly 101 can be used outside of the context of dchpunits. For example, the regulator assembly 101 may be used as bufferbetween a mains power supply and domestic circuitry such as a lightingcircuit. By controlling the voltage waveform, the power consumed can beadjusted to compensate for instantaneous surges in current demand,improve power factor and provide cost savings from improved electricalefficiency, without noticeable effects. Waveform control can also beused to mitigate against fluctuations due to poor quality of supply ordue to peak current demands of high crest factor loads. Indeed, theregulator assembly 101 may be employed to ensure ignition and thenprovide dimming of fluorescent lighting, that cannot be achievedconventionally.

A further application of the regulator assembly 101 according to thepresent invention is to connect a mains supply and an electric motor.The regulator assembly would then operate as a very low cost, simplepower economiser and controller, giving lower motor losses compared tonormal drives.

1. A switching regulator for regulating an ac signal, the regulatorcomprising: input terminals for receiving an input ac signal; outputterminals for providing an output signal; an inductor and a capacitorarranged to smooth the output signal appearing at the output terminals;a positive half-cycle part and a negative half-cycle part arranged toregulate the positive and negative half-cycles respectively of the inputac signal; each of the positive and negative half-cycle partscomprising: a modulating transistor operable to modulate the respectivehalf-cycle of the input ac signal and having an associated modulatordiode arranged to allow current flow through the modulating transistorduring that modulating transistor's respective half-cycle and to resistcurrent flow through the modulating transistor during the otherhalf-cycle; and a clamping diode arranged to protect the modulatingtransistor from reverse-bias voltages and having an associated clampswitch operable to cause the clamping diode to conduct during thatmodulating transistor's respective half-cycle and to cause the clampingdiode to be bypassed during the other half-cycle; the regulator furthercomprising: a first switching controller operable to cause themodulating transistors to switch, comprising a first power supplyoperable to take power from one half-cycle of the input ac signal; and asecond, separate switching controller operable to cause the clampswitches to switch, comprising a second power supply operable to takepower from the other half-cycle of the input ac signal.
 2. The switchingregulator of claim 1, wherein the modulator diodes are arranged toprovide a shunt around their associated modulating transistor.
 3. Theswitching regulator of claim 1, wherein the first switching controlleris operable to cause the modulating transistors to switch to providepulse width modulation of the input ac signal.
 4. The switchingregulator of claim 1, wherein the clamp switches are transistors.
 5. Theswitching regulator of claim 4, wherein the clamp transistors arearranged on respective shunts thereby bypassing, when conducting, theirassociated clamping diode.
 6. The switching regulator of claim 1,wherein the first and second power supplies are arranged to drawsubstantially equal average quantities of power from the input acsignal.
 7. The switching regulator of claim 6, wherein the first andsecond power supplies are arranged to provide substantially differentinstantaneous currents to the first and second switching controllersrespectively.
 8. The switching regulator of claim 7, wherein at leastone of the first and second power supplies decrease the voltage of thesignal drawn from the input ac signal.
 9. The switching regulator ofclaim 8, wherein the at least one power supply comprises a switchoperable to decrease the voltage.
 10. The switching regulator of claim8, wherein the at least one power supply comprises a Zener diodearranged to provide the reduced voltage.
 11. The switching regulator ofclaim 8, wherein at least one of the first or second power suppliescomprises resistors rated to provide a required current.
 12. Theswitching regulator of claim 1, wherein at least one of the first orsecond power supplies comprises smoothing components such as an inductorand/or a capacitor.
 13. The switching regulator of claim 1, wherein atleast one of the first or second power supplies comprises a capacitoroperable to store energy during one half-cycle and to release energyduring the other half-cycle.
 14. The switching regulator of claim 1,wherein the first switching controller and/or the second switchingcontroller has an associated diode arranged to allow only either thepositive or the negative half-cycle of the input ac signal to flow tothe first or second power supply respectively.
 15. The switchingregulator of claim 14, wherein both the first and the second switchingcontrollers have associated diodes that are arranged to be oppositelybiased such that one of the power supplies receives the positivehalf-cycle of the input ac signal and the other of the power suppliesreceives the negative half-cycle of the input ac signal.
 16. Theswitching regulator of claim 1, wherein the first switching controllerincludes a first power supply that is operable to take power from theinput ac signal and supply it to the modulating transistors.
 17. Theswitching regulator of claim 1, wherein the second switching controllerincludes a second power supply that is operable to take power from theinput ac signal and supply it to clamp transistors operating as theclamp switches.
 18. A method of providing power to the switchingregulator of claim 1, comprising: powering the first switchingcontroller by drawing power from one half-cycle of the input ac signal;and powering the second switching controller by drawing power from theother half-cycle of the input ac signal; wherein the average power drawnduring each half-cycle is substantially the same.
 19. The method ofclaim 18, further comprising using the first switching controller toprovide current to the modulating transistors, and using the secondswitching controller to provide current to the clamp switches, whereinthe instantaneous currents provided by the first and second switchingcontrollers are substantially different.
 20. A domestic combined heatand power unit comprising an alternator and the switching regulator ofclaim 1, wherein the switching regulator is arranged to receive thesignal provided by the alternator as the input ac signal.
 21. The unitof claim 20 connected to an electrical grid, wherein the switchingregulator is arranged to provide the output signal to the electricalgrid.
 22. The unit of claim 20 having at least one electrical applianceconnected thereto, wherein the switching regulator is arranged toprovide the output signal to the electrical appliance.
 23. A dimmerswitch for fluorescent lighting comprising any of the switchingregulators of claim 1.